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UC3842BD1 датащи(PDF) 8 Page - Motorola, Inc

номер детали UC3842BD1
подробное описание детали  HIGH PERFORMANCE CURRENT MODE CONTROLLERS
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производитель  MOTOROLA [Motorola, Inc]
домашняя страница  http://www.freescale.com
Logo MOTOROLA - Motorola, Inc

UC3842BD1 датащи(HTML) 8 Page - Motorola, Inc

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UC3842B, 43B UC2842B, 43B
8
MOTOROLA ANALOG IC DEVICE DATA
OPERATING DESCRIPTION
The UC3842B, UC3843B series are high performance,
fixed frequency, current mode controllers. They are
specifically designed for Off–Line and dc–to–dc converter
applications offering the designer a cost–effective solution
with minimal external components. A representative block
diagram is shown in Figure 17.
Oscillator
The oscillator frequency is programmed by the values
selected for the timing components RT and CT. Capacitor CT
is charged from the 5.0 V reference through resistor RT to
approximately 2.8 V and discharged to 1.2 V by an internal
current sink. During the discharge of CT, the oscillator
generates an internal blanking pulse that holds the center
input of the NOR gate high. This causes the Output to be in a
low state, thus producing a controlled amount of output
deadtime. Figure 1 shows RT versus Oscillator Frequency
and Figure 2, Output Deadtime versus Frequency, both for
given values of CT. Note that many values of RT and CT will
give the same oscillator frequency but only one combination
will yield a specific output deadtime at a given frequency. The
oscillator thresholds are temperature compensated to within
±6% at 50 kHz. Also because of industry trends moving the
UC384X into higher and higher frequency applications, the
UC384XB is guaranteed to within
±10% at 250 kHz. These
internal circuit refinements minimize variations of oscillator
frequency and maximum output duty cycle. The results are
shown in Figures 3 and 4.
In many noise–sensitive applications it may be desirable
to frequency–lock the converter to an external system clock.
This can be accomplished by applying a clock signal to the
circuit shown in Figure 20. For reliable locking, the
free–running oscillator frequency should be set about 10%
less than the clock frequency. A method for multi–unit
synchronization is shown in Figure 21. By tailoring the clock
waveform, accurate Output duty cycle clamping can be
achieved.
Error Amplifier
A fully compensated Error Amplifier with access to the
inverting input and output is provided. It features a typical dc
voltage gain of 90 dB, and a unity gain bandwidth of 1.0 MHz
with 57 degrees of phase margin (Figure 7). The
non–inverting input is internally biased at 2.5 V and is not
pinned out. The converter output voltage is typically divided
down and monitored by the inverting input. The maximum
input bias current is –2.0
µA which can cause an output
voltage error that is equal to the product of the input bias
current and the equivalent input divider source resistance.
The Error Amp Output (Pin 1) is provided for external loop
compensation (Figure 31). The output voltage is offset by two
diode drops (
≈1.4 V) and divided by three before it connects
to the non–inverting input of the Current Sense Comparator.
This guarantees that no drive pulses appear at the Output
(Pin 6) when pin 1 is at its lowest state (VOL). This occurs
when the power supply is operating and the load is removed,
or at the beginning of a soft–start interval (Figures 23, 24).
The Error Amp minimum feedback resistance is limited by the
amplifier’s source current (0.5 mA) and the required output
voltage (VOH) to reach the comparator’s 1.0 V clamp level:
Rf(min) ≈
3.0 (1.0 V) + 1.4 V
0.5 mA
= 8800
Current Sense Comparator and PWM Latch
The UC3842B, UC3843B operate as a current mode
controller, whereby output switch conduction is initiated by
the oscillator and terminated when the peak inductor current
reaches the threshold level established by the Error Amplifier
Output/Compensation (Pin 1). Thus the error signal controls
the peak inductor current on a cycle–by–cycle basis. The
Current Sense Comparator PWM Latch configuration used
ensures that only a single pulse appears at the Output during
any given oscillator cycle. The inductor current is converted
to a voltage by inserting the ground–referenced sense
resistor RS in series with the source of output switch Q1. This
voltage is monitored by the Current Sense Input (Pin 3) and
compared to a level derived from the Error Amp Output. The
peak inductor current under normal operating conditions is
controlled by the voltage at pin 1 where:
Ipk =
V(Pin 1) – 1.4 V
3 RS
Abnormal operating conditions occur when the power
supply output is overloaded or if output voltage sensing is
lost. Under these conditions, the Current Sense Comparator
threshold will be internally clamped to 1.0 V. Therefore the
maximum peak switch current is:
Ipk(max) =
1.0 V
RS
When designing a high power switching regulator it
becomes desirable to reduce the internal clamp voltage in
order to keep the power dissipation of RS to a reasonable
level. A simple method to adjust this voltage is shown in
Figure 22. The two external diodes are used to compensate
the internal diodes, yielding a constant clamp voltage over
temperature. Erratic operation due to noise pickup can result
if there is an excessive reduction of the Ipk(max) clamp
voltage.
A narrow spike on the leading edge of the current
waveform can usually be observed and may cause the power
supply to exhibit an instability when the output is lightly
loaded. This spike is due to the power transformer
interwinding capacitance and output rectifier recovery time.
The addition of an RC filter on the Current Sense Input with a
time constant that approximates the spike duration will
usually eliminate the instability (refer to Figure 26).


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